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Module Multilevel-Clamped Composited Multilevel Converter (M-MC2) with Dual T-Type Modules and One Diode Module

  • Luo, Haoze (College of Electrical Engineering, Zhejiang University) ;
  • Dong, Yufei (College of Electrical Engineering, Zhejiang University) ;
  • Li, Wuhua (College of Electrical Engineering, Zhejiang University) ;
  • He, Xiangning (College of Electrical Engineering, Zhejiang University)
  • Received : 2014.03.31
  • Accepted : 2014.10.06
  • Published : 2014.11.20

Abstract

A modular multilevel-clamped composited multilevel converter ($M-MC^2$) is proposed. $M-MC^2$ enables topology reconfiguration, power device reuse, and composited clamping. An advanced five-level converter ($5L-M-MC^2$) is derived from the concept of $M-MC^2$. $5L-M-MC^2$ integrates dual three-level T-type modules and one three-level neutral point clamped module. This converter can also integrate dual three-level T-type modules and one passive diode module by utilizing the device reuse scheme. The operation principle and SPWM modulation are discussed to highlight converter performance. The proposed $M-MC^2$ is comprehensively compared with state-of-the-art five-level converters. Finally, simulations and experimental results are presented to validate the effectiveness of the main contributions of this study.

I. INTRODUCTION

Multilevel converters are generally recognized as reasonable and effective configurations to break down contradictions between limited power device capacities and medium-voltage (MV) high power systems, such as new energy power generation and high power driver systems [1]-[5]. Multilevel converters achieve higher output voltage levels than traditional two-level converters by reducing the voltage steps from indirect power devices in series. Increased output voltage levels result in low dv/dt, which enhances output power quality and decreases insulation damage in MV high power systems.

Voltage source multilevel converters are generally divided into three fundamental groups, namely, diode-clamped converters, flying capacitor (FC) converters, and cascaded H-bridge converters (CHB) [6], [7], which have been successfully applied in industrial areas. Three-level neutral point clamped inverters (3L-NPC) with IGBTs or IGCTs are standard products in MV drives [8]-[10]. Clamped diodes are replaced by active power devices to derive the active 3L-NPC (3L-ANPC) topology and to solve the thermal imbalance in 3L-NPC circuits [11]. However, additional voltage levels are needed to decrease total harmonic distortion (THD) in some high-quality AC systems. The number of diodes or FCs significantly increase when diode-clamped or FC converters are expanded to higher voltagelevel applications. The CHB topology can be easily extended to higher level applications because of its modular design. However, each H bridge cell requires an isolated DC supply, which is generally obtained from multipulse diode rectifiers [12]. The high complexity and cost of phase shifting transformers are the main drawbacks when high voltage levels are required. Advanced multilevel converters are introduced and developed according to combined basic ML topologies to derive high-performance five-level converters with simplified structures. A five-level NPC H-bridge inverter (5L-NPC/H-bridge) is proposed and applied, as shown in Fig. 1(a); these H-bridge cells consist of two three-level NPC circuits [13].

Fig. 1.State-of-the-art five-level topologies: (a) 5L-NPC/H-bridge converter, (b) 5L-Composed topology, (c) 5L-ANPC converter.

Consequently, the power switch numbers are less than those of 5L diode-clamped converters. An additional auxiliary 3L-NPC is inserted into a three-phase 3L-NPC topology to generate a five-level multilevel clamped multilevel converter (5L-MLC2) with fewer power devices [14]. However, the modulation of 5L-MLC2 is slightly complex because the additional auxiliary 3L-NPC unit has three phases. Two half-bridge cells and one 3L-NPC converter are integrated to reduce the number of diodes in 5L diode-clamped converters and to derive a 5L-composed topology, as plotted in Fig. 1(b) [15]. However, the voltage stress of diodes may double in the case of the highest level generation when compared with the switch voltage stress. Furthermore, a five-level hybrid converter composed of active NPC and FC configuration (5L-ANPC) is proposed by ABB and employed in a commercial ACS 2000 system [16], as demonstrated in Fig. 1(c). However, the pre-charging and voltage balance of the FC is a serious concern.

Modular multilevel structures generally increase system reliability and are suitable for large-scale manufacturing. However, most state-of-the-art multilevel topologies are generated only through cell-by-cell combination with different basic units [17]-[19]. Deriving advanced multilevel converters with modular configuration and less power switches remains an attractive research topic. A modular multilevel-clamped composited multilevel converter (M-MC2) is proposed to provide information on modular multilevel topology generation and its advantages, including topology reconfiguration, power device reuse, and composited clamping. On the basis of M-MC2, one 3L-NPC and two 3L-T-type converters are integrated and reconstituted to derive an advanced five-level converter. The derived 5L topology only consists of two three-level T-type modules and one passive diode module because some power switches are shared in the proposed topology.

This paper is organized as follows: In Section II, the proposed concept of M-MC2 is highlighted to derive an advanced five-level converter. In Section III, the operation principle and PWM modulation of the derived 5L-M-MC2 are described. In Section IV, the advantages of the derived converter are explored by comprehensively comparing it with other multilevel converters. In Section V, the simulation and experimental waveforms of the proposed converter are illustrated to demonstrate its superior performance. In Section VI, the conclusions are drawn and summarized.

 

II. MODULAR MULTILEVEL-CLAMPED COMPOSITED MULTILEVEL CONVERTER (M-MC2) CONCEPT

Fig. 2 shows the multilevel-clamped multilevel converters (MLC2) proposed in [14].

Fig. 2.Published MLC2 concept in [14].

The MLC2 solution is different from the conventional multilevel generation law; a set of multilevel clamping units (MCU) is connected to one multilevel output unit (MOU) in the main power converter. MCU is an auxiliary switched circuit unit that conveys additional voltage levels, and the number of auxiliary clamping points depends on the MOU configuration. MCU and MOU could be two-level or multilevel converters. Consequently, the output voltage levels of MOU can be easily expanded by MCU. The topological complexity, which originates from the MLC2 concept, depends on the specific configurations of MCU and MOU. The combined converters generally become more complex as the voltage levels of MCU or MOU increase. However, inner voltage level generation, excluding the outmost voltage levels, should be connected to auxiliary clamping points, as shown in Fig. 2. The power in inner voltage levels should flow through both MCU and MOU. This procedure increases the complexity of the converters derived from MLC2.

The switches in MCU and MOU could be composited to decrease circuit configuration and simplify the power flow path through valuable innovations. Fig. 3 shows an improved modular multilevel-clamped composited multilevel converter (M-MC2) derived from a power device reuse scheme.

Fig. 3.Block diagram of the proposed M-MC2.

The outmost voltage levels and part of the inner voltage levels remain unchanged in the proposed M-MC2, similar to MLC2. However, the rest of the inner voltage levels are directly generated by MOU because of the original function of MOU and the shared power switches in MCU and MOU, which make the auxiliary clamping point dispensable. Consequently, the circuit configuration of derived multilevel converters from M-MC2 can be simplified. An advanced five-level converter (5L-M-MC2) is generated (Fig. 4). This converter is employed to explore the detailed topology generation procedure on the basis of M-MC2.

Fig. 4.Derivative processes of the proposed five-level topology.

Two three-level T-type modules and one 3L-NPC module are integrated and composited. The bidirectional switches in the three-level T-type modules are implemented by two independent or reverse-blocking IGBTs [20]. The series active switches Sa1, Sa02, Sa3, and Sa4 in the two three-level T-type modules coincide with the four series switches in the 3L-NPC module.

According to M-MC2, some power devices could be shared when two structures are combined. Fig. 3 shows that the two 3L-T-type modules serve as MCUs, and the 3L-NPC module acts as an MOU. The outmost voltage levels are directly generated by conducting Sa1, Sa2 or Sa3, Sa4 in the derived 5L-M-MC2. The innermost voltage level is generated by conducting Sa2, Dca1 or Sa3, Dca2, similar to the case of the MOU in the 3L-NPC converter. The other voltage levels are synthesized by generating Sca1/Sca3, Sa2 or Sca2/Sca4, Sa3, which integrates MCUs and the MOU. Consequently, the required power switch number is reduced because the power devices in the dual 3L-T-type and 3L-NPC modules are shared and reused. Both diode- and active-clamped approaches are included in the derived 5L-M-MC2. Key power switches are reused and reconfigured in the MCUs and MOU. Fig. 5 shows that the derived 5L-M-MC2 can easily be extended to three-phase systems.

Fig.5.Circuit configuration of the three-phase 5L-M-MC2 system.

 

III. OPERATION PRINCIPLE AND MODULATION FOR THE DERIVED 5L-M-MC2

A. Switching Pattern

The single phase of the derived 5L-M-MC2 is re-drawn in Fig. 6 for improved switching pattern analysis.

Fig. 6.Single phase of the proposed 5L-M-MC2.

Inverter leg A consists of eight active switches and two clamping diodes. In the following analysis, the voltage across each capacitor is assumed to be E, and the total DC voltage Vd is equally divided by the capacitors (Vd =4E).

Table I shows the operating status of the proposed topology, as represented by the switching states; “1” indicates that the corresponding active switch is turned on, whereas “0” indicates that the switch is turned off. The voltage at terminal A with respect to neutral point Z is the terminal voltage vAZ. Five different output voltage levels are synthesized with the corresponding switching states. Four switches are observed in the conduction states at any time, and the switch pairs [(Sa1, Sca1), (Sa2, Sca2), (Sa3, Sca3), (Sa4, Sca4)] are complementary. Only two of eight active devices in the inverter leg are involved in the transition between adjacent states. The output voltage has five voltage levels, namely, +2E, +1E, 0E, -1E, and -2E.

TABLE IDEFINITION OF SWITCHING STATES

B. Switching Commutation

Fig. 7 illustrates the commutation process with iout > 0. In the switching state “2P,” the load current iout flows though Sa1 and Sa2, whereas Sca3 and Sca4 are gated on. Sca3 and Dca1 are reversed-biased because of the conduction of Sa1. The voltage across the bidirectional switches is equal to E, which indicates that Sca1 is reversed-biased and is thus turned off. When the switching state commutates to state “P,” Sa1 should first be turned off with the current iout naturally commutating through Sca3; Sca1 is then turned on after time δ. The current is diverted from Sca3 to Dca1 when the converter commutates from state “P” to state “O” with iout >0. Sca3 is first turned off, and the path of iout changes from Sca3 to Dca1. Sca4 is reversed-blocked but gated on.

Fig. 7.Commutation between different switching states with iout>0.

Figs. 7(c) to (d) illustrates the other commutations and voltage stress changes. The maximum voltage stress of Sa1 to Sa4 is half of Vd, whereas the two bidirectional switches withstand only a quarter of Vd. The commutation with iout<0 is similar because of the symmetry of the proposed topology.

C. Modulation of the Proposed Converter

Space vector modulation (SVM) is widely used for voltage source inverters because of real-time modulation and high DC bus voltage utilization. However, space vectors and redundant switching states exponentially increase when the number of output voltage levels exceeds three. The location of the reference vector is difficult to determine in the Cartesian coordinate system. However, the fast space vector modulation algorithm based on a hexagonal coordinate system avoids several trigonometric operations and reduces computational workload [21]. This fast algorithm is applicable to the proposed 5L-M-MC2. Level-shifted multicarrier modulation is also extensively employed in multilevel converter applications. This modulation is divided into three schemes: in-phase disposition (IPD), alternative phase opposite disposition (APOD), and phase opposite disposition (POD). The IPD modulation scheme provides a notable harmonic profile, which is discussed in [22]. Fig.8 presents the key waveforms of a five-level inverter modulated by the IPD carrier-based modulation scheme.

Fig. 8.Simulation waveforms in the proposed converter using IPD modulation (fo=50Hz, fsw=1kHz, M=0.95).

Four carrier waves (vcar1, vcar2, vcar3, and vcar4) are disposed vertically. Gate signals (vg1, vg2, vg3, and vg4) for Sa1, Sa2, Sa3, and Sa4in Fig. 8 are generated at the intersections of the carrier waveforms and the modulation wave. The gate signals of Sca1, Sca2, Sca3, and Sca4 are complementary to vg1, vg2, vg3, and vg4 and are thus, not plotted in Fig. 8. The modulation of the proposed 5L-M-MC2 is straightforward and easily implemented.

 

IV. SIGNIFICANT CONVERTER FEATURES AND PERFORMANCE COMPARISON

The practical establishment of the proposed 5L-M-MC2 is simpler than that of conventional multilevel topologies because only three modules are required (i.e., two 3L-T-type modules and one diode module). The modularized structure facilitates the bus–bar design and modular production in high-power applications. The clamping diodes and active switches could withstand different voltage stresses. For instance, when the switching state “2P” commutates to state “P” with iout > 0, the voltage across switches Sa3 and Sa4 re 2E and 3E/2, respectively. However, the maximum voltage stresses of the power devices could withstand half of the total DC voltage during commutation, whereas the two bidirectional switches could withstand only one quarter of Vd. The bidirectional switches consist of low-voltage devices, which significantly reduces the cost of 5L-M-MC2.

Table II presents a comparison of the state-of-the-art 5L topologies. The conventional 5L diode-clamped converter requires eight switches and 12 diodes with Vdc/4 voltage rating. The number of diodes is significantly reduced to two in 5L-M-MC2. Unlike the conventional 5L FC converter, 5L-M-MC2 does not require bulky and costly fly capacitors. Although the 5L-CHB circuit requires a minimum component quantity, the six necessary isolated DC supplies increase the system cost and converter size in the three-phase system.

TABLE IICOMPARISON OF STATE-OF-THE-ART FIVE-LEVEL TOPOLOGIES

The three-phase 5L-M-MC2 system only requires one DC source. Although the ANPC-FC topology does not require clamping diodes, the voltage balance of the FC requires an additional control scheme. Assuming that the bidirectional switches are implemented by RB-IGBTs, the load current only flows through two power devices during the switching states. However, a minimum of three power devices are involved in the flowing path in other topologies. M-MC2 features are optimized by combining various basic multilevel clamping units.

 

V. SIMULATION AND EXPERIMENTAL RESULTS

To verify the validity of the proposed converter, a three-phase system is simulated in Matlab/Simulink. The simulated waveforms of the three-phase inverter are illustrated in Fig. 9. The DC input voltage is supplied by four independent 1000 V voltage sources, whereas the entire DC bus voltage reaches 4000 V. All capacitors have the same capacitance (1000 μF). A three-phase series R-L load is connected to the AC side with Ro=100 Ω and Lo=10mH. The fundamental frequency is set to 50 Hz, whereas the modulation frequency is 5 kHz. The modulation index (M) is set to 0.95. Fig. 10 shows the line-to-line voltage vAB in 40 ms.

Fig. 9.Line-to-line voltage vAB for 5L-M-MC2 (M=0.95).

Fig.10.Voltage stress of the main devices in the single phase leg.

The line-to-line voltage is close to the sinusoidal waveform because it has nine voltage levels. Fig. 10 shows the voltage stress of the main power devices. The voltage rating of Sa1 and Sa2 is Vd/2 or Vd/4. The maximum blocking voltage for active switches Sa1 and Sa2 is half of Vd .The voltage stress of the bidirectional switches is Vd/4 or Vd/8, whereas the maximum voltage stress is only one quarter of Vd. The output line-to-line voltage decreases from nine levels to three levels when the modulation index in the proposed 5L-M-MC2 changes from 1 to 0. Fig. 11 shows the total THD and output voltage level for line-to-line voltage with different values of M.

Fig. 11.Simulated THD of line-to-line voltage with modulation index.

A prototype of the three-phase converter is built and tested with the proposed topology. The T-type 3-L module is F3L80R12W1H3_B11 from Infineon, and the clamping diode is the SKKD_75F12 diode module from SEMIKRON. The prototype is controlled by a digital signal processing board from TI (TMS320F28335). The DC input voltage is supplied by four independent voltage sources of 17 V, and each series capacitor has the same capacitance of 1410 μF. The output frequency is set as 50 Hz, whereas the modulation frequency is 5 kHz. The dead time is set to 2 μs. Fig. 12 shows the experimental results of the single-phase voltage vAZ for M = 0.8. The phase output voltage has five levels. Fig. 13(a) shows that the measured line-to-line voltage vAB has seven output levels for M = 0.8. Fig. 1(b) shows that the line-to-line voltage vAB contains nine levels for M = 0.95.

Fig.12.Experimental results of phase voltage vAZ with fo=50Hz.

Fig. 13.Experimental results of line-to-line voltage vAB for fo = 50 Hz: (a) voltage waveforms of vAB in M = 0.8, (b) voltage waveforms of vAB in M = 0.95.

Fig. 14 shows the measured voltage stress of the main power devices. Figs. 14(a) to 14(c) show the voltage waveforms of switches Sa1, Sa2, and Dca1, respectively. The maximum voltage stress of the main power devices is half of the dc voltage, which is 2E. The main devices have the same voltage stresses as NPC-3L. Fig. 14(d) shows that the maximum value of vXU across Sca1/Sca1 is Vd/4, which implies that the bidirectional switches are achieved by low-voltage power devices. The experimental results are consistent with the simulated results in Figs. 12 to 14. Consequently, the simulated and experimental results confirm the validity of the proposed topology.

Fig. 14.Experimental results of main power devices: (a) voltage waveforms of switch Sa1, (b) voltage waveforms of switch Sa2., (c) voltage waveforms of diode Dca1, (d) voltage waveforms of bidirectional switch Sca1/Sca3.

 

VI. CONCLUSIONS

This study proposes 5L-M-MC2 for multilevel converters. A streamlined topology can be generated though the organic composition of two or more basic units by reusing power devices. An advanced 5L-M-MC2 with a simplified structure is derived from two 3L-T type modules and one diode module. The operational principle and PWM modulation are analyzed in detail. The proposed 5L-M-MC2 is compared with other state-of-the-art topologies to present its advantages. Simulation and experimental results verify the feasibility of the derived 5L-M-MC2. Future works could focus on the balancing of capacitor voltage and its future applications.

Acknowledgement

Supported by : National Basic Research Program of China, National Nature Science Foundations of China

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